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SP6127
High-Voltage, Step-Down Controller in TSOT6
FEATURES Wide 4.5V - 29V Input Voltage Range Internal compensation Built-in High-Current PMOS Driver Adjustable Overcurrent Protection Internal soft start 900kHz Constant Frequency Operation 0.6V Reference Voltage 1% output setpoint accuracy Lead Free, RoHS Compliant Package: Small 6-pin TSOT
LX
GND
FB
6
5
4
SP6127
6 PinTSOT
1
VIN
2
GATE
3
VDR
__________________________________________________________ DESCRIPTION
The SP6127 is a PWM controlled step down (buck) voltage mode regulator with VIN feedforward and internal Type-II compensation. It operates from 4.5V to 29V VIN, making it suitable for 5V, 12V and 24V applications. By using a PMOS driver, this device is capable of operating at 100% duty cycle. The highside driver is designed to drive the gate 5V below VIN. The programmable overcurrent protection is based on the high-side MOSFET's ON resistance sensing and allows setting the overcurrent protection value up to 300mV threshold (measured between VIN-LX). The SP6127 is available in a space-saving 6pin TSOT package making it the smallest controller available capable of operating from 24VDC supplies.
_______________________________________ TYPICAL APPLICATION CIRCUIT
VIN
Q1 Si2343DS 2 1 Vin Rs=1k LX C7 0.1uF 6 L1, IHLP-2525CZ 3.3uH, 30mOhm, 6A Gate C1 4.7uF
12V
GND VOUT
C4 22uF R1 200k, 1% 4 CZ 33pF R2 100k, 1% D1 1N4148
SP6127
3 VDR
Ds MBRA340T3G
RZ 2K
1.8V 0-2.0A
VFB GND 5 SHDN High=Of f
GND
Jan28-08 RevE
SP6127 TSOT-6 PFET Buck Controller Page 1
2007 Exar Corporation
ABSOLUTE MAXIMUM RATINGS
These are stress ratings only, and functional operation of the device at these ratings or any other above those indicated in the operation sections of the specifications below is not implied. Exposure to absolute maximum rating conditions for extended periods of time may affect reliability. Input Voltage....................................................-0.3V to 30V Lx.............................................................-2V to 30V FB.................................................................-0.3V to 5.5V Storage Temperature.............................-65 C to 150 C Junction Temperature...................................-40C to 125C Lead Temperature (Soldering, 10 sec)...................300 C ESD Rating..................1kV LX, 2kV all other nodes, HBM
____________________________________________ ELECTRICAL SPECIFICATIONS
Specifications are for TAMB=TJ=25C, and those denoted by apply over the full operating range, -40C< Tj <125C. Unless otherwise specified: VIN =4.5V to 29V, CIN = 4.7F.
PARAMETER UVLO Turn-On Threshold UVLO Turn-Off Threshold UVLO Hysteresis Operating Input Voltage Range Operating Input Voltage Range Operating VCC Current Reference Voltage Accuracy Reference Voltage Accuracy Reference Voltage Reference Voltage Switching Frequency Peak-to-peak ramp Voltage Minimum ON-Pulse Duration Minimum Duty Cycle Maximum Duty Cycle Gate Driver Turn-Off Resistance Gate Driver Pull-Down Resistance Gate Driver Pull-up Resistance VIN - VDR voltage difference Overcurrent Threshold LX pin Input Current OFF interval during hiccup Soft start time SHDN Threshold SHDN Threshold Hysteresis
MIN 4.2 4.0
TYP 4.35 4.15 0.2
MAX 4.5 4.4
UNITS V V V
CONDITIONS
0C< Tj <125C 0C< Tj <125C 0C< Tj <125C
4.5 7 0.3 0.5 0.5 0.594 0.588 750 0.6 0.6 900 VIN/5 40 100 50 4 3 4.5 270 25 3 0.8 300 30 70 5 1.0 100
29 29 3 1 2 0.606 0.612 1050 100 0 60 8 6 5.5 330 35 9 1.2
V V mA % % V V kHz V ns % % k
VFB=1.2V

Internal resistor between GATE and VIN VIN=12V, VFB=0.5V, Measure resistance between GATE and VDR VIN=12V, VFB=0.7V, Measure resistance between GATE and VIN
V mV uA ms ms V mV
Measure VIN - VDR, VIN>7V Measure VIN - LX VLX = VIN VFB=0.58V, measure between VIN=4.5V and first GATE pulse
Apply voltage to FB
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SP6127 TSOT-6 PFET Buck Controller Page 2
2007 Exar Corporation
_______________________________________________________ PIN DESCRIPTION
PIN # 1 2 3 PIN NAME VIN GATE VDR DESCRIPTION Input power supply for the controller. Place input decoupling capacitor as close as possible to this pin. Connect to the gate terminal of the external P-channel MOSFET. Power supply for the internal driver. This voltage is internally regulated to about 5V below VIN. Place a 0.1F decoupling capacitor between VDR and VIN as close as possible to the IC. Regulator feedback input. Connect to a resistive voltage-divider network to set the output voltage. This pin can be also used for ON/OFF control. If this pin is pulled above 1V the P-channel driver is disabled and controller resets internal soft start circuit. Ground pin. This pin is used as a current limit input for the internal current limit comparator. Connect to the drain pin of the external MOSFET through an optional resistor. Internal threshold is pre-set to 300mV nominal and can be decreased by changing the external resistor based on the following formula: VTRSHLD = 300mV - 30uA * R
4 5
FB GND
6
LX
______________________________________________________ BLOCK DIAGRAM
5V VIN
VDR Oscillator Vin - 5V LDO
VIN 5V Internal LDO I = k x VIN
FAULT VREF
PWM Latch Reset Dominant S GATE
FB
+ + -
R
Error Amplifier
PWM Comparator VDR FAULT
FAULT ENBL UVLO FAULT Register S 4-Bit counter
-
LX
70ms delay
Overcurrent Comparator VIN - 0.3V
+ -
R R Set Dominant
POR
1V
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SP6127 TSOT-6 PFET Buck Controller Page 3
+
30uA
GND
2007 Exar Corporation
____________________________________________________ GENERAL OVERVIEW
The SP6127 is a fixed frequency, Voltage-mode, non-synchronous PWM controller optimized for minimum component, small form factor and cost effectiveness. It has been designed for singlesupply operation ranging from 4.5V to 29V. SP6127 has Type-II internal compensation for use with Electrolytic/Tantalum output capacitors. For ceramic capacitors Type-III compensation can be implemented by simply adding an R and C between output and Feedback. A precision 0.6V reference, present on the positive terminal of the Error Amplifier, permits programming of the output voltage down to 0.6V via the FB pin. The output of the Error Amplifier is internally compared to a feed-forward (VIN/5 peak-to-peak) ramp and generates the PWM control. Timing is governed by an internal oscillator that sets the PWM frequency at 900kHz. SP6127 contains useful protection features. Overcurrent protection is based on the high-side MOSFET's RDS(ON) and is programmable via a resistor placed at LX node. Under-Voltage LockOut (UVLO) ensures that the controller starts functioning only when sufficient voltage exists for powering IC's internal circuitry. SP6127 Loop Compensation The SP6127 includes Type-II internal compensation components for loop compensation. External compensation components are not required for systems with tantalum or aluminum electrolytic output capacitors with sufficiently high ESR. Use the condition below as a guideline to determine whether or not the internal compensation is sufficient for your design.
where:
f ESRZERO = f DBPOLE =
1 2. .R ESR .C OUT
........... (2)
1 2. . L C OUT
............ (3)
Creating a Type-III compensation Network The above condition requires the ESR zero to be at a lower frequency than the double-pole from the LC filter. If this condition is not met, Type-III compensation should be used and can be accomplished by placing a series RC combination in parallel with R1 as shown below. The value of CZ can be calculated as follows and RZ selected from table 1.
CZ =
L C .............. (4) 1.25 x R1
fESRZERO/ fDBPOLE
1X 2X 3X 5X >= 10X
RZ
50K 40K 30K 10K 2K
Type-II internal compensation is sufficient if the following condition is met:
f ESRZERO < f DBPOLE ................... (1)
Table1- Selection of RZ
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2007 Exar Corporation
______________________________________________________ GENERAL OVERVIEW
SP6127
Vout
CP1
2pF RZ
CZ2
130pF
RZ2
200k VFB
CZ
R1 200k, 1%
Error Amplif ier
Figure 1- RZ and CZ in conjunction with internal compensation components form a TypeIII compensation network Loop Compensation Example 1- A converter utilizing a SP6127 has a 3.3H inductor and a 22F/5m ceramic capacitor. Determine whether Type-III compensation is needed. From equation (2) response of the circuit, seen in figure 2, validates the above procedure. Loop Compensation Example 2- A converter utilizing the SP6127 has a 3.3H inductor and a 220F, 82m Aluminum Electrolytic capacitor. Determine whether Type-III compensation is needed. From equation (2)
fESRZERO
= 1.45MHz. From
equation (3) fDBPOLE = 18.4 kHz. Since the condition specified in (1) is not met, Type-III compensation must be used by adding external components RZ and CZ. Using equation (4) CZ is calculated to be 34pF (use 33pF). Following the guideline given in table 1, a 2k RZ should be used. The steps followed in example 1 were used to compensate the typical application circuit shown on page 1. Satisfactory frequency
Figure 2- Satisfactory frequency response of typical application circuit shown on page 1. Crossover frequency fc is 100kHz with a corresponding phase margin of 60 degrees.
Jan28-08 RevE SP6127 TSOT-6 PFET Buck Controller Page 5 2007 Exar Corporation
+ Vref =0.6V R2
fESRZERO
= 8.8kHz. From
equation (3) fDBPOLE = 5.9kHz. Since the condition specified in (1) is not met, Type-III compensation needs to be used by adding external components RZ and CZ. Using equation (4) CZ is calculated 108pF (use 100 pF). Since fESRZERO /fDBPOLE is approximately 2, RZ must be set at 40k .
______________________________________________________ GENERAL OVERVIEW
Overcurrent Protection (OCP)
Vin
SP6127
Gate Q1
for Rs in the range of 0.5k to 3k . For Rs larger than 3k , test results are lower than those predicted by (5), due to circuit parasitics. Therefore the maximum value of Rs should be limited to 3k . Note that in order to safeguard against false overcurrent trigger due to transients, there is a 150ns delay between the turn on of the MOSFET and when OCP circuit is activated. As a consequence at very high Vo/VIN ratio, where MOSFET on-time is less than 150ns, the OCP circuit will not detect overcurrent.
Ov er-Current Comparator LX Rs
Figure 3 Overcurrent protection circuit
The overcurrent protection (OCP) circuit functions by monitoring the voltage across the high-side FET Q1. When this voltage exceeds 0.3V, the overcurrent comparator triggers and the controller enters hiccup mode. For example if Q1 has RDS(ON)=0.1 , then the overcurrent will trigger at I = 0.3V/0.1 =3A. To program a lower overcurrent use a resistor Rs as shown in figure 1. Calculate Rs from:
Rs =
Where:
0.3 - (1.15 x Iout x Rds (on) x Kt ) ...(5) 30uA
1.15 is used to calculate peak inductor current which is nominally 15% higher than average output current RDS(ON) is MOSFET ON-resistance rating Kt is a multiplier that accounts for increase in RDS(ON) due to temperature Example: A switching MOSFET used with SP6127 has RDS(ON) of 0.08 and Kt is 1.5. Program the over-current circuit so that maximum output is 2A.
Rs =
Rs=800
0.3 - (1.15 x 2 A x 0.08Ohm x 1.5) 30uA
Using the above equation there is good agreement between calculated and test results
Jan28-08 RevE
+ 30uA Vin - 0.3V
Ds
Using the ON/OFF Function The Feedback pin serves a dual role of ON/OFF control. The MOSFET driver is disabled when a voltage greater than 1V is applied at the FB pin. Maximum voltage rating of this pin is 5.5V. The controlling signal should be applied through a small signal diode as shown on page 1. Please note that an optional 10k bleeding resistor across the output helps keep the output capacitor discharged under no load condition. Programming the Output Voltage To program the output voltage, calculate R2 using the following equation:
R2 =
R1 Vout Vref - 1
Where: VREF=0.6 is the reference voltage of the SP6127 R1=200k is a fixed-value resistor that, in addition to being a voltage divider, it is part of the compensation network. In order to simplify compensation calculations, R1 is fixed at 200k .
Soft Start Soft Start is preset internally to 5ms (nominal). Internal Soft Start eliminates the need for the external capacitor CSS that is commonly used to program this function.
SP6127 TSOT-6 PFET Buck Controller Page 6
2007 Exar Corporation
______________________________________________________ GENERAL OVERVIEW
MOSFET Gate Drive The P-channel drive is derived through an internal regulator that generates VIN-5V. This pin (VDR) must be connected to VIN with a 0.1F decoupling capacitor. The gate drive circuit swings between VIN and VIN-5 and employs powerful drivers for efficient switching of the Pchannel MOSFET. Power MOSFET Selection Select the Power MOSFET for Voltage rating BVDSS, On resistance RDS(ON), and thermal resistance RTHJA. BVDSS should be about twice as high as VIN in order to guard against switching transients. The recommended MOSFET voltage rating for VIN of 5V, 12V and 24V is 12V, 30V and 40V respectively. RDS(ON) must be selected such that when operating at peak current and junction temperature, the Overcurrent threshold of the SP6127 is not exceeded. Allowing 50% for temperature coefficient of RDS(ON) and 15% for inductor current ripple, the following expression can be used: voltage selection. For a low duty cycle application such as the circuit shown on first page, the Schottky diode is conducting most of the time and its conduction losses are the largest component of losses in the converter. Conduction losses can be estimated from:
Vout Pc = Vf x Iout x 1 - Vin
where: VF is diode forward voltage at IOUT The Schottky diode's AC losses due to its switching capacitance are negligible. Inductor Selection Select the Inductor for inductance L and saturation current ISAT. Select an inductor with ISAT higher than the programmed overcurrent. Calculate inductance from:
0.3V RDS (ON ) 1.5 x 1.15 x Iout
Within this constraint, selecting MOSFETs with lower RDS(ON) will reduce conduction losses at the expense of increased switching losses. As a rule of thumb, select the highest RDS(ON) MOSFET that meets the above criteria. Switching losses can be assumed to roughly equal to the conduction losses. A simplified expression for conduction losses is given by:
Vout 1 1 L = (Vin - Vout ) x x x Vin f Irip
where: VIN is converter input voltage VOUT is converter output voltage f is switching frequency IRIP is inductor peak-to-peak current ripple (nominally set to 30% of IOUT) Keep in mind that a higher IRIP results in a smaller inductor which has the advantages of small size, low DC equivalent resistance DCR, high saturation current ISAT and allows the use of a lower output capacitance to meet a given step load transient. A higher IRIP, however, increases the output voltage ripple and increases the current at which converter enters Discontinuous Conduction Mode. The output current at which converter enters DCM is 1/2 of IRIP. Note that a negative current step load that drives the converter into DCM will result in a large output voltage transient. Therefore the lowest current for a step load should be larger than 1/2 of IRIP.
Vout Pcond = Iout 2 x RDS (ON ) x Vin
The MOSFET's junction temperature can be estimated from:
T = (2 x Pc x Rthja ) + Tambient
Schottky Rectifier selection Select the Schottky Diode for Voltage rating VR, Forward voltage Vf, and thermal resistance RTHJA. The Voltage rating should be selected using the same guidelines outlined for MOSFET
Jan28-08 RevE
SP6127 TSOT-6 PFET Buck Controller Page 7
2007 Exar Corporation
______________________________________________________ GENERAL OVERVIEW
Output Capacitor Selection Note that a smaller inductor results in a higher Select the output capacitor for voltage rating, capacitance and Equivalent Series Resistance (ESR). Nominally the voltage rating is selected to be twice as large as the output voltage. Select the capacitance to satisfy the specification for output voltage overshoot/undershoot caused by current step load. A steady-state output current IOUT corresponds to inductor stored energy of 1/2 L IOUT2. A sudden decrease in IOUT forces the energy surplus in L to be absorbed by COUT. This causes an overshoot in output voltage that is corrected by the reduced duty cycle of the power switch. Use the following equation to calculate
IRIP, therefore requiring a larger COUT and/or lower ESR in order to meet VRIP.
Input Capacitor Selection Select the input capacitor for Voltage, Capacitance, ripple current, ESR and ESL. Voltage rating is nominally selected to be twice the input voltage. The RMS value of the input capacitor current, assuming a low inductor ripple current (IRIP), can be calculated from:
COUT:
Icin = Iout x D(1 - D )
In general, total input voltage ripple should be kept below 1.5% of VIN (not to exceed 180mV). Input voltage ripple has three components: ESR and ESL cause a step voltage drop upon turn on of the MOSFET. During on time, the capacitor discharges linearly as it supplies IOUT-IIN. The contribution to Input voltage ripple by each term can be calculated from:
I 2 2 - I12 Cout = L x Vos 2 - Vout 2
Where: L is the output inductance
I2 is the step load high current I1 is the step load low current
Vos is output voltage including overshoot VOUT is steady state output voltage Output voltage undershoot calculation is more complicated. Test results for SP6127 buck circuits show that undershoot is approximately equal to overshoot. Therefore the above equation provides a satisfactory method for calculating COUT. Select ESR such that output voltage ripple (VRIP) specification is met. There are two components to VRIP: The first component arises from charge transferred to and from COUT during each cycle. The second component of VRIP is due to inductor ripple current flowing through the output capacitor's ESR. It can be calculated from:
V , Cin =
Iout x Vout x (Vin - Vout ) fs x Cin x Vin 2
V , ESR = ESR(Iout - 0.5Irip )
V , ESL = ESL
(Iout - 0.5Irip )
Trise
Where TRISE is the rise time of current through capacitor Total input voltage ripple is sum of the above:
V , Tot = V , Cin + V , ESR + V , ESL
In circuits where converter input voltage is applied via a mechanical switch, excessive ringing may be present at turn-on that may interfere with smooth startup of the SP6127. The addition of an inexpensive 100F Aluminum Electrolytic capacitor at the input will help reduce ringing and restore a smooth startup.
1 Vrip = Irip x ESR + 8 x Cout x fs
2
2
Where: IRIP is inductor ripple current fs is switching frequency COUT is output capacitor calculated above
Jan28-08 RevE
SP6127 TSOT-6 PFET Buck Controller Page 8
2007 Exar Corporation
_____________________________________ TYPICAL PERFORMANCE CHARACTERISTICS
VIN
Q1 Si2343DS 2 1 Vin Rs=1k LX C7 0.1uF 6 L1, IHLP-2525CZ 3.3uH, 30mOhm, 6A Gate C1 4.7uF
12V
GND VOUT
C4 22uF R1 200k, 1% 4 CZ 33pF R2 100k, 1% D1 1N4148
SP6127
3 VDR
Ds MBRA340T3G
RZ 2K
1.8V 0-2.0A
VFB GND 5 SHDN High=Of f
GND
Figure 4- Application circuit
SP6127 Efficiency versus Iout, Vin=12V,Ta=25C
90
Vout=1.8V
SP6127 Load Regulation Vin=12V
1.810
Efficiency (%)
80
Vout (V)
1.805
70
1.800
60
1.795
50 0.0 0.5 1.0 Iout (A) 1.5 2.0
1.790 0.0 0.5 1.0 Iout (A) 1.5 2.0
Figure 5- Efficiency, natural convection
Figure 6- Load regulation
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SP6127 TSOT-6 PFET Buck Controller Page 9
2007 Exar Corporation
_____________________________________ TYPICAL PERFORMANCE CHARACTERISTICS
Figure 7- Step load 1-2A ch1: VIN ch2: VOUT, ch3: IOUT
Figure 8- Overcurrent shutdown ch1: VIN, ch2: VOUT, ch3: Inductor current, ch4: IOUT
Figure 9- Startup no load ch1: VIN, ch2: VOUT, ch3: IOUT
Figure 10- Start up 2A ch1: VIN ch2: VOUT, ch3: IOUT
Figure 11- Output ripple at 0A is 32mV ch1: VIN, ch2: VOUT, ch3: IOUT
Figure 12- Output ripple at 2A is 12mV ch1: VIN, ch2: VOUT, ch3: IOUT
Jan28-08 RevE
SP6127 TSOT-6 PFET Buck Controller Page 10
2007 Exar Corporation
______________________________________________________ PACKAGE: 6PIN TSOT
Jan28-08 RevE
EXAR
SP6127 TSOT-6 PFET Buck Controller Page 11 2007 Exar Corporation
ORDERING INFORMATION Part Number Temperature Range Package SP6127EK1-L..................................-40C to +125C...........................(Lead Free) 6 Pin TSOT SP6127EK1-L/TR..............................-40C to +125C...........................(Lead Free) 6 Pin TSOT
/TR = Tape and Reel Pack Quantity for Tape and Reel is 2500
For further assistance: Email: EXAR Technical Documentation: customersupport@exar.com http://www.exar.com/TechDoc/default.aspx? Exar Corporation Headquarters and Sales Office 48720 Kato Road Fremont, CA 94538 main: 510-668-7000 fax: 510-668-7030 EXAR Corporation reserves the right to make changes to the products contained in this publication in order to improve design, performance or reliability. EXAR Corporation assumes no responsibility for the use of any circuits described herein, conveys no license under any patent or other right, and makes no representation that the circuits are free of patent infringement. Charts and schedules contained here in are only for illustration purposes and may vary depending upon a user's specific application. While the information in this publication has been carefully checked; no responsibility, however, is assumed for inaccuracies. EXAR Corporation does not recommend the use of any of its products in life support applications where the failure or malfunction of the product can reasonably be expected to cause failure of the life support system or to significantly affect its safety or effectiveness. Products are not authorized for use in such applications unless EXAR Corporation receives, in writing, assurances to its satisfaction that: (a) the risk of injury or damage has been minimized; (b) the user assumes all such risks; (c) potential liability of EXAR Corporation is adequately protected under the circumstances.
Jan28-08 RevE
SP6127 TSOT-6 PFET Buck Controller Page 12
2007 Exar Corporation

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